Method and apparatus for implementing a channel correction in a digital data link

ABSTRACT

The present invention concerns a method and apparatus for implementing channel equalization on a digital communications path. According to the method, the outgoing bit stream is coded into symbols, the channel distortion is compensated for by means of symbol preceding (TML), the precoded symbols are sent over a communications channel ( 2,  CHN), whereby means are provided for recovering symbols that have passed over the communications channel ( 2,  CHN) and via the signal processing means of the receiver into a bit stream. According to the invention, during the data transmission state, the preceding (TML) is adjusted based on the content values of the precoder delay line and the value of such an error variable (e) that is dependent on the difference between the signal measurable at the receiver and the value of the symbol sent (S) by the transmitter or estimated (S′) by the receiver, in which receiver the signal is measured at a point where said difference attains the minimum value of its absolute value when the adjustment is in a steady state.

[0001] The invention relates to a method according to the preamble ofclaim 1 for implementing channel equalization on a digitalcommunications path.

[0002] The invention also relates to an apparatus according to claim 17suited for performing channel equalization on a digital communicationspath, as well as a transmitter according to claim 22 and a receiveraccording to claim 23.

[0003] In the transmission of digital data, or a bit stream, over acommunications channel 2, the bit stream is converted in a transmitter(TX) into an analog signal that is capable of passing through thecommunications channel. The communications channel may be a radio path,copper wireline or fiber-optic cable. On the basis of the receivedanalog signal, the receiver (RX) performs a recovery of the sent bitstream as error-free as possible. The bit stream reconstructionperformed in the receiver is complicated by signal distortion and noisesummed with the signal on the communications channel. Due to theseside-effects, a portion of the reconstructed bits are erroneous (e.g.,on an average, 1 bit per 10⁷ bits may be erroneous).

[0004] The signal distortion originating from the transmission path isgenerally compensated for by means of equalizers that are located in thereceiver, the transmitter or partially in both of these. The equalizersmay be of a fixed or adaptive type. Respectively, the effect of noise iscompensated for by means of different coding techniques such asReed-Solomon coding, convolution coding, trellis coding, turbo codingand others.

[0005] A generally used correction method of channel distortion is theuse of a linear adaptive equalizer (FFE). However, a linear equalizeralone may give an insufficient correction on certain channels. This kindof situation may be encountered when the transfer function of the signalband includes zero points, whereby certain frequency components cannotbe passed over the communications channel 2. Then, a feedback equalizeris used to compensate for the distortion caused by the spectral nulls ofthe signal band. Also in a system wherein the channel 2 has no spectralnulls, the use of a feedback equalizer is often advantageous inasmuch itimproves the noise tolerance of the system. If the feedback equalizer islocated in the receiver, it is called a decision-feedback equalizer(DFE), while an equalizer located in the transmitter is called aTomlinson-Harashima precoder. A system may also have both a DFE and aTML. Furthermore, the linear equalizer may be situated in the receiver,the transmitter or a portion of the equalizer may be in the transmitterwhile the other portion is in the receiver.

[0006] In the text describing the prior art and the features of thepresent invention, the following abbreviations are used: CAP Carrierlessamplitude and phase modulation DFE Decision-feedback equalizer FFEFeedforward equalizer, also known as a linear equalizer PAM Pulseamplitude modulation QAM Quadrature amplitude modulation RX Receiver TXTransmitter TML Tomlinson-Harashima precoder.

[0007] In the following, a digital communications channel is examined interms of the training phase of its adaptive equalizers. The line codeused on the channel may be implemented using either pulse-amplitudemodulation (PAM), quadrature-amplitude modulation (QAM) or carrierlessamplitude and phase modulation (CAP). In FIG. 1 is shown a model for asystem implemented using conventional techniques, wherein the receiveris provided with an adaptive linear equalizer (FFE) and an adaptivedecision-feedback equalizer (DFE) (cf. Lee & Messerschmitt). The effectof fixed filters and possible modulation schemes are included in thechannel noise model (CHN). The outgoing bit stream is coded into symbols(S) that are sent through the channel 2. In the receiver, the outputsignal of the channel 2 is processed by equalizers (FFE and DFE), andthe decisions on symbols (S′) are made from the equalized signal. Thedecision resulting in the resolved symbol (S′) is also called theestimated received symbol. Both adaptive equalizers are adapted to thecharacteristics of the channel 2 during the training period carried outwhen a connection is being established. The equalizers are alsocontinually adjusted during the period of data transmission in order tocompensate for possible changes in the channel 2. The equalizers areadapted and controlled on the basis of the detection error (e) of thereceive signal.

[0008] In FIG. 2 is shown another system according to the prior art (cf.Lee & Messerschmitt). The receiver has an adaptive linear equalizer(FFE), while the transmitter has a feedback equalizer (of the TML type).During the teaching period, also this system operates in the samefashion as that illustrated in FIG. 1 using a linear equalizer and adecision-feedback equalizer (DFE). At the end of the training period,the tap-weight values of the decision-feedback equalizer (DFE) aretransmitted over an upstream auxiliary channel to the transmitter,wherein they are utilized in the configuration of a Tomlinson-Harashimaprecoder (TML). The linear equalizer (FFE) of the receiver is adjustedduring the data transmission state, but due to the fixed configurationof the decision-feedback equalizer (TML) of the receiver, the latterequalizer will not be adjusted.

[0009] A benefit of Tomlinson-Harashima preceding over a DFE is thatprecoding does not cause feedback of a detection error as is the case ina DFE. Particularly when the shape of the amplitude response of thecommunications channel 2 is such that large values of tap coefficientsmust be used in the DFE, a really complex problem evokes from thefeedback of erroneous decision-making in the detector. In the mostserious situations, a single erroneous decision may cause loss ofconnection when in a system using a DFE.

[0010] Generally, changes in the characteristics of a communicationschannel 2 can be compensated for by adjusting the linear equalizeralone. However, in some cases the communications channel 2 may includeanalog bandstop filters serving to eliminate radio-frequencyinterference, for instance. The positions of the spectral nulls causedby the analog bandstop filters in the frequency spectrum may vary as thecomponent values of the filters change with temperature. This kind ofvariation in the characteristics of the communications channel 2 cannotbe compensated for simply by adjusting the linear equalizer. Anothercomplication arises from the incapacity of the system to cope in anoptimal manner with varying noise conditions if the decision-feedbackequalizer is not adjusted during the data transmission state.

[0011] In FIG. 3 is illustrated a prior-art method used for solving theabove-described problem. Herein, the system comprises a linear equalizer(FFE), a Tomlinson-Harashima precoder (TML) and decision-feedbackequalizer (DFE). During the training period, the system comprises theFFE and the DFE alone. At the end of the training period, the tapcoefficient values of the DFE are sent to the precoder (TML) included inthe transmitter and the tap coefficient values of the DFE are reset tozero. During the data transmission state, both the FFE and the DFE areadjusted, but not the precoder (TML). A benefit of this arrangement isthat the problems associated with such changes in the communicationschannel characteristics and noise conditions that cannot be coped withmerely by adjusting the linear equalizer are overcome, because also theDFE of the receiver can be adjusted during the data transmission state.A disadvantage still remains from the risk of erroneous decisionfeedback due to the DFE of the receiver. The tap coefficients of the DFEin the receiver may be assumed to have smaller values than in thesituation illustrated in FIG. 1 inasmuch a portion of the feedbackequalization is performed already in the transmitter. Consequently, alsothe effect of erroneous decision feedback is less severe than in theconfiguration shown in FIG. 1. However, the system performance remainssubstantially dependent on how large changes may occur in thecharacteristics of the communications channel 2 and system noisecondition in regard to the preceding situation prevailed during thetraining period.

[0012] A straightforward approach to improve the system shown in FIG. 2or 3 would be to compute the incremental values of tap coefficientadjustments in the receiver from the detector error and symbol decisionsin the same manner as when adjusting a DFE, but then transmitting thecomputed incremental values of adjustment over an auxiliary channel ofthe reverse transmit direction to the transmitter. These incrementaladjustment values are then used for updating the tap coefficient valuesof the precoder in the transmitter. Accordingly, the precoder could beadjusted also during the data transmission state, whereby the receiverDFE would become redundant or the high values of its tap coefficientscan be limited. However, it can be shown that this kind of equalizeradjustment method is not practicable in a general case.

[0013] It is an object of the present invention to overcome thedrawbacks of the above-described techniques and to provide an entirelynovel type of method and apparatus for use on a digital communicationschannel.

[0014] The goal of the invention is achieved by way of adjusting aconventional Tomlinson-Harashima precoder during the data transmissionstate. The adjustment is implemented in a system according to theinvention so that the detector input signal of the receiver istransmitted back to the transmitter over an auxiliary channel of thereverse transmit direction and the tap coefficients of the precoder arethen adjusted on the basis of the detector error and with the help ofthe precoder delay line using, e.g., an LMS algorithm.

[0015] More specifically, the method according to the invention ischaracterized by what is stated in the characterizing part of claim 1.

[0016] Furthermore, the apparatus according to the invention ischaracterized by what is stated in the characterizing part of claim 17.

[0017] The transmitter according to the invention is characterized bywhat is stated in the characterizing part of claim 22.

[0018] The receiver according to the invention is characterized by whatis stated in the characterizing part of claim 23.

[0019] The invention offers significant benefits.

[0020] The invention allows the precoder to adapt in a continuous manneralso during the data transmission state to changes occurring in thenoise conditions on the communications channel 2 and also to changes inthe properties of analog filters due to temperature variations and todrift caused by other factors.

[0021] In the following, the invention is described in more detail withreference to exemplifying embodiments elucidated in the appendeddrawings in which

[0022]FIG. 1 shows a block diagram of a system of the prior art forimplementing channel equalization;

[0023]FIG. 2 shows a block diagram of a second system of the prior artfor implementing channel equalization;

[0024]FIG. 3 shows a block diagram of a third system of the prior artfor implementing channel equalization;

[0025]FIG. 4 shows a simplified block diagram of a system according tothe invention;

[0026]FIG. 5 shows a more detailed block diagram of a system accordingto the invention and its mathematical model;

[0027]FIG. 6 shows a block diagram of a system according to theinvention not having equalizers at its receive end;

[0028]FIG. 7 shows a block diagram of a system according to theinvention having merely a linear equalizers at its receive end;

[0029]FIG. 8 shows a block diagram of a system according to theinvention having only a decision-feedback equalizer at its receive end;

[0030]FIG. 9 shows a block diagram of a system according to theinvention having both a linear equalizer and a decision-feedbackequalizer;

[0031]FIG. 10 shows a block diagram of a system according to theinvention that has the linear equalizer divided into two separateequalizers in a cascaded configuration and further includes adecision-feedback equalizer; and

[0032]FIG. 11 shows a block diagram of a system according to theinvention that has the linear equalizer paralleled with an adaptivefilter (that is, a second linear equalizer) and further includes adecision-feedback equalizer.

[0033] Each one of the equalizers in FIGS. 1-11 may be adjustable orfixed except for the precoder TML that according to the invention in allcases is adjustable.

[0034] Accordingly, the invention concerns a method and apparatussuitable for implementing a communications system, wherein aTomlinson-Harashima precoder is adjusted during the data transmissionstate, see FIG. 4. The theoretical basis of the method will be evidentto the reader from the discussion given below.

[0035] Referring to FIG. 5, therein is shown a discrete-time model of asystem equipped with Tomlinson-Harashima precoding. The result of themodulo operation, which is an integral part of the precoding step, isincluded in the transmitted symbol. Basics on Tomlinson-Harashimaprecoding can be found, e.g., in cited reference (Lee & Messerschmitt).

[0036] The following symbol notification will be used in the discussion:

[0037] C(z⁻¹) transfer function of communications channel 2 (includesfixed filters, modulation systems, etc.)

[0038] E(z⁻¹) transfer function of linear equalizer

[0039] S_(k) kth sent symbol of extended symbol constellation=initialsymbol constellation (S_(k))+result of modulo operation (m_(k))

[0040] S′_(k) symbol decision for above sent symbol using extendedsymbol constellation in receiver

[0041] d_(k) detector input signal in receiver for above sent symbol

[0042] e_(k) detector error in symbol decision S′_(k)

[0043] h₀, h₁, h₂, . . . impulse responses for H(z⁻¹); H(z⁻¹)=C(z⁻¹)E(z⁻¹)

[0044] v₀, v₁, v₂, . . . V_(n) tap coefficients of Tomlinson-Harashimaprecoder

[0045] b_(k) output signal of Tomlinson-Harashima precoder

[0046] Referring to FIG. 5, the precoder output is:$b_{k} = {S_{k} - {\sum\limits_{i = 1}^{n}{b_{k - i}v_{i}}}}$

[0047] In the system response function, the precursor equalizer sets tozero all the tap coefficients preceding the decision-making tap (maintap), whereby the impulse response of

[0048] C(z⁻¹)E(z⁻¹)=H(z⁻¹) is of the form:

[0049] 0, 0, . . . , 0, 1, r_(k), r_(k+1), r_(k+2), . . .

[0050] To simplify the notation, the zero-set taps are ignored(corresponding to an ideal delay) and the impulse response of H(z⁻¹) isdenoted as follows:

[0051] 1, h₁, h₂, . . . , where h₁=r_(k), h₂=r_(k+1), . . .

[0052] Respectively, the detector input is:$d_{k} = {b_{k} + {\sum\limits_{i = 1}^{\infty}{b_{k - i}h_{i}}}}$a  n  d${d_{k} = {{S_{k} - {\sum\limits_{i = 1}^{n}{b_{k - i}v_{i}}} + {\sum\limits_{i = 1}^{\infty}{b_{k - i}h_{i}}}} = {S_{k} + {\sum\limits_{i = 1}^{n}{b_{k - i}\left( {h_{i} - v_{i}} \right)}} + {\sum\limits_{i = {n + 1}}^{\infty}{b_{k - i}h_{i}}}}}},$

[0053] wherefrom the value of error variable is obtained:$u_{k} = {{d_{k} - S_{k}} = {{\sum\limits_{i = 1}^{n}{b_{k - i}\left( {h_{i} - v_{i}} \right)}} + {\sum\limits_{i = {n + 1}}^{\infty}{b_{k - i}{h_{i}.}}}}}$

[0054] The thus obtained error u_(k) is equal to the detector error(e_(k)) measured at the receiver if a correct symbol decision (that is,S′_(k)=S_(k)) has been made.

[0055] It can be shown that the successive values of Tomlinson-Harashimaprecoder outputs (b_(k), b_(k+1), b_(k+2), . . . ) are noncorrelatingwith each other (cf. Lee & Messerschmitt). Thence, a parameter valuethat is required in the adjustment of the precoder tap coefficients andis proportional to the tap coefficient error (h_(i)−v_(i)) can beobtained by correlating the detector error with an element of theprecoder delay line, that is:

E{b _(k-1) *u _(k)}=σ_(b) ²(h _(i) −v ₁),

since

E{b _(k-i) *b _(k-i)}=σ_(b) ² (σ_(b) ²=output power of precoder) and E{b_(k-i) *b _(k-j)}=0, when i≠j.

[0056] Herefrom, it is possible in principle to solve the values of tapcoefficients with which the precoder can most precisely compensate forthe distortion in the communications channel 2. Solving the equationsgives:${v_{i}^{n\quad e\quad w} = {{v_{i} + \frac{E\left\{ {b_{k - i}^{*}u_{k}} \right\}}{\sigma_{b}^{2}}} = h_{i}}},$

[0057] where v₁ ^(new) is the updated tap coefficient of the precoderand index i=1 . . . n (number of taps). In the equations, superindexnotation “*” refers to a complex conjugate.

[0058] From the above analysis, it is obvious that the tap coefficientsof a Tomlinson-Harashima precoder can be adjusted by the least meansquares algorithm (LMS) using the error difference between the detectorinput and the transmitted symbol (dk−Sk), and the values contained bythe precoder delay line. In a practicable implementation, the formula ofthe estimated values is replaced by the following control algorithm oftap coefficients:

v ₁ ^(uew) =v _(i) +μb _(k-i) *u _(k),

[0059] where μ is the control increment. The equation may also bewritten using the value e_(k) of the error.

[0060] By way of a similar analysis, a variable which is proportional tothe error in the adjustment of the tap coefficients of the DFE locatedin the receiver can be written as:

E{S′ _(k-i) *e _(k)}=σ_(S) ²(h ₁ −q _(i)),

[0061] where q_(i) is the DFE tap coefficient and σ_(S) ² is the powerof the received symbols. The symbol decision (S′) of the extended symbolconstellation must herein be replaced by the symbol decision of theoriginal symbol constellation if the transmitter is not provided with aprecoder. The detector error (e_(k)) is equal to u_(k) if a correctsymbol decision (S′_(k-i)=S_(k-i)) has been made. Comparison of theabove equations reveals that the tap coefficient adjustment terms of theprecoder and the DFE are equal if the symbol decisions are equal to theoutput values of the precoder. Symbol decisions, except for erroneoussymbol decisions, are equal to the output values of the precoder if theprecoder tap coefficient values are zero. When the values of theprecoder tap coefficients are increased, also the difference (power ofdifference) between the symbol decisions and the output values of theprecoder increases.

[0062] By simulation, it can be shown that the adjustment of theprecoder with the help of the control algorithm of the DFE starting fromthe zero-reset taps is stable as long as the values of the precoder tapcoefficients remain sufficiently small. Herein, it must be admitted thatthe term “sufficiently small” is difficult to define in an exact mannerinasmuch each case is individually subject to strongly varyingconstraints. If large values of precoder tap coefficients are requiredfor the compensation of channel distortion, the system operation becomesunstable when the control loop has increased the precoder tapcoefficients to so high values that the differences between the symboldecisions and the precoder output values grow excessively large.

[0063] A practical problem in the method according to the inventionarises from the requirement of a correct mutual phase between thereceiver input signal values (d_(k)) and the values of the elements(b_(k−1), b_(k−2), b_(k−3), . . . ) of the precoder delay line. In anactual situation, this detail can be handled by synchronizing theinformation on the error variable to the precoder delay line contentelements with the help of the line frame synchronization information.Additionally, a functional apparatus needs memory elements for storageof the symbol and delay line content elements until the moment when theinformation related to their respective error variable has beensubmitted to the transmitter.

[0064] The method according to the invention for adjusting the tapcoefficients of a Tomlinson-Harashima precoder in the data transmissionstate is accomplished as follows:

[0065] 1. The error variable (u_(k) or e_(k)) to be used in theadjustment of the precoder is defined as the difference (d_(k)−S_(k) ord_(k)−S′_(k)) between the signal (d_(k)) detected at receiver and thesent symbol (S) or, respectively, the estimated symbol (S′). In thereceiver, the signal d_(k) to be taken at a point where said differencereaches the minimum value of its absolute value when the precoderadjustment is in a steady state.

[0066] 2. The information (d_(k) or e_(k)) used in the determination ofthe error variable value is transmitted from the receiver to thetransmitter over an auxiliary channel of the reverse transmit direction.−3. The precoder tap coefficients are adjusted with the help of an LMSalgorithm on the basis of the values of the error variable (d_(k)−S_(k)or e_(k)=d_(k)−S′_(k)) and the precoder delay line content elements(b_(k−1), b_(k−2), b_(k−3), . . . ).

[0067] Systems according to the invention using the precoder adjustmentbased on the control scheme of the invention applied during the datatransmission state is illustrated in FIGS. 6, 7, 8, 9, 10 and 11. Thesystems shown therein comprise a Tomlinson-Harashima precoder (TML) thatis adjusted by means of the method according to the invention. Thecomputation of the error variable value (subtraction operation) neededfor the adjustment of the precoder can be made in either the transmitteror the receiver.

[0068] In the system shown in FIG. 6, only a precoder is used forimplementing the channel equalization. The precoder is adjusted based onthe error (e₁) that represents the situation preceding detection. Hence,the system illustrated in FIG. 6 is functional only on such channelsthat do not cause precursor intersymbol interference (precursor ISI),that is, an interaction between successive symbols prior to the instantof decision-making.

[0069] The system shown in FIG. 7 has both a precoder and a linearequalizer in the receiver. The precoder and the linear equalizer areadjusted based on the error (e₁) corresponding to the situationpreceding detection.

[0070] The system shown in FIG. 8 has a precoder and a decision-feedbackequalizer (DFE) in the receiver. The precoder is adjusted based on theerror (e₁) corresponding to the situation before the receive signal iscorrected by the effect of the DFE. The DFE is adjusted in aconventional manner based on the value of the error e₂. The system shownin FIG. 8 is functional only on such channels that do not causeprecursor ISI.

[0071] The communications of the error variable information between thereceiver and the transmitter and the need for synchronization betweenthe transmitted information and the values of the precoder delay linecontent elements make this kind of precoder adjustment slow in practicaloperation. Generally, such slow response is not a major problem, becausethe goal of the equalizer adjustment during the data transmission stateis to respond to changes in the characteristics of the communicationschannel 2 that, due to temperature variations, occur at a slow rate.Hence, the system illustrated in FIG. 7 is suited for a major number ofpractical situations. However, there may also appear needs for a fastadjustment of the decision-feedback equalizer. An example of such a caserepresents a situation where a narrowband interference falling on thesignal transmission spectrum must be eliminated by a stop band (spectralnull) created in the linear equalizer. While the stop band generated bythe linear equalizer manages to eliminate the interference, it alsocauses in the data signal a distortion that must be compensated for byadapting the decision-feedback equalizer to the new situation. In thiskind of a case, the adjustment rate of the decision-feedback equalizermust be in the same order with the adjustment rate of the linearequalizer.

[0072] Embodiments according to the invention serving to solve theabove-described problems are illustrated in FIGS. 9, 10 and 11.

[0073] The system shown in FIG. 9 has a precoder and a linear equalizer(FFE) in the receiver, complemented with a decision-feedback equalizer(DFE). The precoder is adjusted based on the value of the error (e₁) inthe signal immediately after the linear equalizer. The linear equalizerand the DFE are adjusted in a conventional manner based on the value ofthe error e₂.

[0074] Next, a situation is discussed wherein fast adaptation ofequalization is needed and adaptation of the linear equalizer alone isinsufficient. The linear equalizer and the DFE have a fast response andhence are quickly adapted to the changed situation. As to the precoder,the situation is similar to a system configuration having no DFE,because e₁=e₂ if the output of the DFE is zero. Resultingly, theprecoder adapts at its inherent adaptation rate so as to make the errore₁ smaller. The linear equalizer and the DFE detect the adaptation ofthe precoder in the same manner as any slow change in the communicationschannel 2 and adjust themselves accordingly. The effect of post cursorISI is eliminated from the output signal of the linear equalizer as soonas the precoder has reached a new steady state. Hereupon, the tapcoefficients of the DFE should be (almost) zero and, in a practicalsituation, the tap coefficients have been adjusted to zero or a lowvalue during the adaptation period of the precoder.

[0075] The method according to the invention makes it possible toconfigure a system wherein the DFE of the receiver can respond to needsof fast adaptation, but still the compensation settings of the DFE forchannel distortion can be reflected to the transmitter at the speeddetermined by the adjustment rate of the precoder. This arrangementlimits the increase of tap coefficients in the DFE and thus reduces thefeedback of receive error.

[0076] In the system shown in FIG. 9, the DFE is involved with theadjustment process of the precoder, because the DFE affects the error e₂on the basis of which the linear equalizer is adjusted that in turnaffects the error e₁. Resultingly, the system stability is deterioratedat certain mutual adjustment rates of the precoder, the linear equalizerand the DFE. Hence, the system illustrated in FIG. 9 requires carefuldesign in respect to these adjustment rates.

[0077] A situation, wherein the DFE does not affect the adjustment ofthe precoder, can be accomplished by using systems illustrated in FIGS.10 and 11. In the system shown in FIG. 10, the linear equalizer isdivided in two cascaded separate linear equalizers (FFE1 and FFE2). Theprecoder and the first linear equalizer (FFE1) are adjusted based on theerror (e₁) measurable at the output of the first linear equalizer. Thesecond linear equalizer (FFE2) and the DFE are adjusted in aconventional manner based on error e₂. During operation, the precoderseeks determined by its inherent adjustment rate toward a situationwherein no feedback equalization is needed in the receiver. Thisarrangement avoids the use of high tap coefficient values in the DFE.The sampling rates of the cascaded adaptive filters FFE1 and FFE2 can beequal or different. Furthermore, the sampling rates can be equal to thesymbol rate, multiples thereof or rational number multiples thereof.According to a preferred embodiment, the sampling rate of FFE1 is amultiple of the symbol rate (fractional spaced equalizer) and thesampling rate of FFE2 is equal to the symbol rate. According to anotherpreferred embodiment, the sampling rates of both filters FFE1 and FFE2can be equal to or multiples of the symbol rate.

[0078] In the system shown in FIG. 11, the linear equalizer (FFE) isparalleled by another adaptive FIR filter (AFIR) that also is a linearequalizer. The precoder and the adaptive filter (AFIR) are adjustedbased on the error (e₁) measurable at the output of the adaptive filter.The linear equalizer (FFE) and the DFE are adjusted in a conventionalmanner based on error e₂. Also in this system, the precoder seeksdetermined by its inherent adjustment rate toward a situation wherein nofeedback equalization is needed in the receiver, thus avoiding high tapcoefficient values in the DFE. An advantage of this arrangement is thatthe adaptive filter AFIR needs adjustment and coefficient computationonly when the precoder is being adjusted. Hence, the AFIR filter can beimplemented computationally without the need for complicated ASICdesign.

[0079] In the above-described systems illustrated in FIG. 6 . . . 11,the linear equalizer and/or the decision-feedback equalizer (DFE) of thereceiver may in certain cases be such that needs no adjustment duringthe data transmission state. Furthermore, the linear equalizer and/orthe DFE may comprise an entirely fixed filter configuration, wherebythis filter is not adjusted even during the training period. Obviously,the replacement of an adaptive equalizer by an entirely fixedconfiguration or a configuration which is not adjustable during thetraining period compromises the system capability of adapting to changesin the communications channel parameters.

REFERENCES

[0080] [Lee & Messerschmitt] E. A. Lee and D. G. Messerschmitt, DigitalCommunication, Kluwer Academic Publishers, 1994.

what is claimed is:
 1. Method for implementing channel equalization on adigital communications path, the method comprising the steps of codingthe outgoing bit stream into symbols, compensating for channeldistortion by means of symbol precoding (TML), sending the precodedsymbols over a communications channel (2, CHN), whereby recoveringsymbols that have passed over the communications channel (2, CH) and viathe signal processing means of the receiver into a bit stream,characterized in that during the data transmission state, the precoding(TML) is adjusted based on the content values of the precoder delay lineand the value of such an error variable (e) that is dependent on thedifference between the signal measurable at the receiver and the sent(S) or estimated (S′) symbol value, in which receiver the signal ismeasured at a point where said difference attains the minimum value ofits absolute value when the adjustment is in a steady state.
 2. Methodaccording to claim 1, characterized in that the value of the errorvariable is computed at the receive end (3).
 3. Method according toclaim 1, characterized in that the value of the error variable iscomputed at the trait end (1).
 4. Method according to claim 1, 2 or 3,characterized in that the channel distortion is compensated for at thereceive end (3) by means of a linear equalizer (FFE), whose outputsignal is processed to obtain the value of an error variable (e₁) thatis used for adjusting both said precoder (TML) and said linear equalizer(FFE).
 5. Method according to claim 1, 2 or 3, characterized in that thechannel distortion is compensated for at the receive end (3) by means ofa linear equalizer (FFE) and a decision-feedback equalizer (DFE) andthat the value of an error variable (e₁) associated with the outputsignal of the linear equalizer (FFE) is used for adjusting the precoder(TML) and that the value of another error variable (e₂) associated withthe input signal of the receiver detector is used for adjusting thelinear equalizer (FFE) and the decision-feedback equalizer (DFE). 6.Method according to claim 1, 2 or 3, characterized in that the channeldistortion is compensated for at the receive end (3) by means ofcascaded linear equalizers (FFE1 and FFE2) and a decision-feedbackequalizer (DFE) and that the value of an error variable (e₁) associatedwith the output signal of the first linear equalizer (FFE1) is used foradjusting the precoder (TML) and said first linear equalizer (FFE1) andthat the value of another error variable (e₂) associated with the inputsignal of the receiver detector is used for adjusting the second linearequalizer (FFE2) and the decision-feedback equalizer (DFE).
 7. Methodaccording to claim 1, 2 or 3, characterized in that the channeldistortion is compensated for at the receive end (3) by means ofparalleled linear equalizers (AFIR and FFE) and a decision-feedbackequalizer (DFE) and that the value of an error variable (e₁) associatedwith the output signal of one linear equalizer (AFIR) is used foradjusting the precoder (TML) and said first linear equalizer (AFIR) andthat the output of the second linear equalizer (FFE) is coupled to theentity formed by the receiver detector and the decision-feedbackequalizer (DFE) and that the value of another error variable (e₂)associated with the input signal of the receiver detector is used foradjusting said second linear equalizer (FFE) and the decision-feedbackequalizer (DFE).
 8. Method according to any one of the foregoing claims,characterized in that the tap coefficients of the precoder (TML) areadjusted with the help of a least-mean-squares algorithm (LMS) based onthe content values of the precoder delay line and using the errorvariable (e₁) mentioned in any of the foregoing claims as the errorvariable.
 9. Method according to claim 8, characterized in that the tapcoefficients (v_(i)) values of the precoder (TML) are adjusted using atap coefficient control algorithm written as v_(i)^(new)=v_(i)+μb_(k-i)*u_(k), where v_(i) ^(new) is the updated value ofa tap coefficient, v_(i) is the value of the tap coefficient prior tothe update operation, b_(k-i)* is the conjugate value of the content ofthe precoder delay line and, u_(k) is the value of the error variableand μ is the control step.
 10. Method according to any one of theforegoing claims, characterized in that the information that is relatedto the error variable and is to be transmitted from the receiver overthe reverse direction auxiliary channel to the transmitter for adjustingthe precoder located therein is synchronized to values of the precoderdelay line content elements with the help of the line framesynchronization information.
 11. Method according to claim 6,characterized in that the sampling rates of the linear equalizers (FFE1and FFE2) are equal to a multiple or rational number multiple of thesymbol rate and that said sampling rates are either equal or different.12. Method according to claim 6, characterized in that the sampling rateof one equalizer (FFE1) is a multiple of the symbol rate, while thesampling rate of the other equalizer (FFE2) is equal to the symbol rate.13. Method according to claim 6, characterized in that the samplingrates of both filters (FFE1 and FFE2) are equal to the symbol rate. 14.Method according to claim 7, characterized in that the sampling rates ofboth linear equalizers (AFIR and FFE) are equal to a multiple orrational number multiple of the symbol rate and that said sampling ratesare either equal or different.
 15. Method according to claim 7,characterized in that the sampling rate of one equalizer (FFE) is amultiple of the symbol rate, while the sampling rate of the otherequalizer (AFIR) is equal to the symbol rate.
 16. Method according toclaim 7, characterized in that the sampling rates of both filters (AFIRand FFE) are equal to the symbol rate.
 17. Apparatus for implementingchannel equalization on a digital communications path, the apparatusbeing formed by a transmitter that further comprises means forconverting a bit stream into symbols, a precoder (TML) for precoding thesymbols, a communications channel (2, CHN) over which the precodedsymbols can be transmitted, and a receiver with signal processing meanssuited for recovering symbols which have passed over the communicationschannel (2, CHN) into a bit stream, characterized in that said apparatuscomprises a precoder (TML) that during the data transmission state canbe adjusted based on the content values of the precoder delay line andthe value of such an error variable (e) that is dependent on thedifference between the signal measurable at the receiver and the sent(S) or estimated (S′) symbol value, in which receiver the signal ismeasured at a point where said difference attains its minimum absolutevalue when the adjustment is in a steady state, and means fordetermining said error variable (e).
 18. Apparatus according to claim17, characterized in that the apparatus at its receive end (3) includesa linear equalizer (FFE) whose output signal is processable to obtainthe value of an error variable (e₁) that is used for adjusting both saidprecoder (TML) and said linear equalizer (FFE).
 19. Apparatus accordingto claim 17, characterized in that the apparatus at its receive end (3)includes a linear equalizer (FFE) and a decision-feedback equalizer(DFE) and that the value of an error variable (e₁) associated with theoutput signal of the linear equalizer (FFE) is usable for adjusting theprecoder (TML) and that the value of another error variable (e₂)associated with the input signal of the receiver detector is usable foradjusting the linear equalizer (FFE) and the decision-feedback equalizer(DFE).
 20. Apparatus according to claim 17, characterized in that theapparatus at its receive end (3) includes cascaded linear equalizers(FFE1 and FFE2) and a decision-feedback equalizer (DFE) and that thevalue of an error variable (e₁) associated with the output signal of thefirst linear equalizer (FFE1) is usable for adjusting the precoder (TML)and the first linear equalizer (FFE1) and that the value of anothererror variable (e₂) associated with the input signal of the receiverdetector is usable for adjusting the second linear equalizer (FFE2) andthe decision-feedback equalizer (DFE).
 21. Apparatus according to claim17, characterized in that the apparatus at its receive end (3) includesparalleled linear equalizers (AFIR and FFE) and a decision-feedbackequalizer (DFE) so that the value of an error variable (e₁) associatedwith the output signal of the first linear equalizer (AFIR) is usablefor adjusting the precoder (TML) and said one linear equalizer (AFIR)and that the output of the second linear equalizer (FFE) is coupled tothe entity formed by the receiver detector and the decision-feedbackequalizer (DFE) and that the value of another error variable (e₂)associated with the input signal of the receiver detector is usable foradjusting said second linear equalizer (FFE) and the decision-feedbackequalizer (DFE).
 22. Transmitter (1) for implementing channelequalization in the method according to claim 1 on a digitalcommunications path, the transmitter (1) including means for convertinga bit stream into symbols, and a precoder (TML) for precoding thesymbols, characterized in that said transmitter (1) comprises means forreceiving such information wherefrom is processable an error variable(e) that is dependent on the difference between the signal measurable atthe receiver and the sent (S) or estimated (S′) symbol value, and aprecoder (TML) that during the data transmission state is adjustablebased on the content values of the precoder delay line and the value ofsaid error variable.
 23. Receiver (3) for implementing channelequalization in the method according to claim 1 on a digitalcommunications path, the receiver (3) including means for convertingsymbols into a bit stream, characterized in that said receiver (3)comprises means for sending to the transmitter of the method over areverse direction auxiliary channel such information wherefrom isprocessable an error variable (e) that is dependent on the differencebetween the signal measurable at the receiver and the sent (S) orestimated (S′) symbol value.
 24. Apparatus according to claim 23,characterized in that the apparatus includes a linear equalizer (FFE)whose output signal is processable to obtain the value of an errorvariable (e₁) that is used for adjusting both said precoder (TML) andsaid linear equalizer (FFE).
 25. Apparatus according to claim 23,characterized in that the apparatus includes a linear equalizer (FFE)and a decision-feedback equalizer (DFE) and that the value of an errorvariable (e₁) associated with the output signal of the linear equalizer(FFE) is usable for adjusting the precoder (TML) and that the value ofanother error variable (e₂) associated with the input signal of thereceiver detector is usable for adjusting the linear equalizer (FFE) andthe decision-feedback equalizer (DFE).
 26. Apparatus according to claim23, characterized in that the apparatus includes cascaded linearequalizers (FFE1 and FFE2) and a decision-feedback equalizer (DFE) andthat the value of an error variable (e₁) associated with the outputsignal of the first linear equalizer (FFE1) is usable for adjusting theprecoder (TML) and the first linear equalizer (FFE1) and that the valueof another error variable (e₂) associated with the input signal of thereceiver detector is usable for adjusting the second linear equalizer(FFE2) and the decision-feedback equalizer (DFE).
 27. Apparatusaccording to claim 23, characterized in that the apparatus includesparalleled linear equalizers (AFIR and FFE) and a decision-feedbackequalizer (DFE) so that the value of an error variable (e₁) associatedwith the output signal of the first linear equalizer (AFIR) is usablefor adjusting the precoder (TML) and said one linear equalizer (AFIR)and that the output of the second linear equalizer (FFE) is coupled tothe entity formed by the receiver detector and the decision-feedbackequalizer (DFE) and that the value of another error variable (e₂)associated with the input signal of the receiver detector is usable foradjusting said second linear equalizer (FFE) and the decision-feedbackequalizer (DFE).